The present invention relates to switching power converters with zero-voltage switching of the power transistors of a resonant-transition power converter.
The power converter portion of modern electronic equipment tends to be bulky and is often the limiting factor when attempting to miniaturize. In reducing power converter size, designers have turned to increased switching frequencies. Higher frequencies allow for smaller, lighter inductive and capacitive energy storage devices, but also bring with them increased switching power losses.
The power dissipation of DC-DC power converters can be reduced by using zero-voltage switching techniques. Zero-voltage switching occurs when a power device begins conduction with a near zero-voltage across the device. Achieving zero-voltage switching over a large range of line voltages and loads is desirable to reduce electromagnetic interference. Previous approaches to achieve zero-voltage switching over large line voltage and load variations has increased the conduction losses of the power devices by increasing the currents carried by the switching devices after they were turned on, resulting in little or no overall gain in efficiency.
A practical example of zero-voltage switching occurs in the power devices of a phase-shifted full bridge converter under certain operating conditions. In the phase-shifted full bridge converter, shown in FIG. 1, MOSFET switches Q1, Q2, Q3, and Q4 operate at a fixed frequency, and the on time of diagonally conducting power devices is not varied as in pwm bridge, but rather the power devices in each leg (one inverter leg having Q1 and Q2, the other leg having Q3 and Q4) are made to alternately conduct at a duty cycle approaching 50%, as can be seen in the waveform diagrams shown in FIG. 2. The phase shift between the operation of the devices of each of the legs determines when diagonal switches are conducting at the same time and therefore suppling power to a load. By varying the phase shift, the resulting output voltage can be pulse width modulated. In the converter of FIG. 1, the transformer primary current current flowing at turn-off of one transistor charges the parasitic capacitances of that transistor while reducing the charge on the parasitic capacitances of the other transistor in the same leg, thereby reducing the voltage across the other transistor, which is also the next transistor to be turned on. As a condition of zero-voltage switching, the turn-on of the transistor in the same leg with the transistor that was just turned off, must be delayed until the voltage across the transistor has been reduced to near zero. For a pair of transistors in the same leg, the time required to charge the capacitances of the transistor being turned off and discharge the parasitic capacitances of the transistor to be turned on, is inversely proportional to the square of the magnitude of current established before the switching interval.
Assuming negligible magnetizing current, the transformer primary current established before the switching interval is different for the two inverter legs in the phase shifted, full bridge inverter. This occurs since Q3 and Q4 are turned on after diagonally conducting transistors (either Q2 and Q3, or Q1 and Q4) where delivering power to the load, and the transformer primary current established before the switching interval is the reflected output inductor current. Transistors Q1 and Q2 are turned on after a freewheeling period when current was being circulated in the bridge, and the current established before switching is the current circulating in the transformer primary during the freewheeling interval. The freewheeling interval is the portion of each cycle when no energy is being supplied to the output from the input power source. The transformer leakage inductance is the energy source displacing charge on the parasitic capacitances of transistors Q1 and Q2, where the magnitude of energy is proportional to the square of the circulating current. The circulating current will decay during the freewheeling interval, as a result of both output rectifiers, which are connected to the two ends of the transformer secondary, conducting current and reducing the energy stored in the leakage inductance. The circulating current is equal to the difference in output rectifier currents divided by the turns ratio of the transformer. With both output rectifiers carrying equal current, the transformer primary current would be zero, resulting in no energy available in the transformer leakage inductance to charge the parasitic capacitances of the transistor to be turned on, and for this condition, zero-voltage switching is not achieved. The magnitude of the circulating current is always less than the reflected output inductor current when power is being delivered to the load. Therefore, zero-voltage switching is more difficult to achieve with transistors Q1 and Q2, which are turned on after circulating current was flowing in the bridge.
One approach to maintain zero-voltage switching is to increase the magnitude of the leakage inductance. This will reduce the magnitude of the current decay during the freewheeling interval. Additional leakage inductance will also increase the energy available for displacing charge on the transistor's output capacitances. With this approach, a minimum leakage inductance can be specified to meet zero-voltage switching requirements for a specific line and load condition. However, the high leakage inductance will reduce the effective duty cycle ratio of the transformer secondary due to the increased recovery time of the output diodes. This will limit the input voltage range of the converter and adversely affect the voltage control characteristics.
Another approach for achieving zero-voltage switching over a wide input line and output load range uses saturable reactors with specific blocking characteristics. A saturable reactor is used in series with each push-pull output rectifier. This technique uses a round or flat B-H loop reactor core material to induce a significant flux excursion during the output rectifier commutating interval. The reactor will provide a blocking characteristic proportional to the flux excursion. The blocking characteristic of the reactor together with a clamped primary prevents the conduction of both rectifiers during the freewheeling interval. This forces the circulating inductor to follow the output inductor current. Therefore, more energy is available to displace the transistor's parasitic capacitance charge. However, for a converter to achieve zero-voltage switching at a light load, the saturable core must be designed to block the entire freewheeling interval. At full load, more than the required energy for zero-voltage switching is available, and as a consequence, transistor conduction losses are increased.
It is an object of the present invention to provide zero-voltage switching with reduced conduction losses in resonant-transition DC-DC converters.
It is another object of the present invention to provide zero-voltage switching with reduced conduction losses in a resonant-transition DC-DC converter over both large line voltage changes and load changes.
It is yet another object of the present invention to provide zero-voltage switching with reduced conduction losses in a resonant-transition DC-DC converter over large line voltage changes and load changes without reducing the input voltage range or adversely affecting the voltage control characteristics of the converter.